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Operazionnie ysiliteli ,ZAP/AZP & (ïðîäîëæåíèå)
milstar: 1941: First (vacuum tube) op-amp An op-amp, defined as a general-purpose, DC-coupled, high gain, inverting feedback amplifier, is first found in US Patent 2,401,779 "Summing Amplifier" filed by Karl D. Swartzel Jr. of Bell labs in 1941. This design used three vacuum tubes to achieve a gain of 90dB and operated on voltage rails of ±350V. ###################################################### It had a single inverting input rather than differential inverting and non-inverting inputs, as are common in today's op-amps. Throughout World War II, Swartzel's design proved its value by being liberally used in the M9 artillery director designed at Bell Labs. ######################################################################### This artillery director worked with the SCR584 radar system to achieve extraordinary hit rates (near 90%) that ####################################################################### would not have been possible otherwise.[3] ########################### http://en.wikipedia.org/wiki/Operational_amplifier
milstar: y down converting and centering the band of interest at DC. The conversion is done by multiplying the input data with a quadrature sinusoid. https://www.renesas.com/eu/en/www/doc/datasheet/hsp50016.pdf
milstar: DDCs convert a real, time domain signal into a complex one, centered at baseband. The process of frequency conversion is achieved by mixing – or multiplying – the input signal with a digital sinusoid at the center of the bandwidth of interest. This creates copies of the signal of interest centered around zero, and also at twice the sinusoid frequency. http://www.ni.com/example/31525/en/
milstar: The NCO, sometimes called a local oscillator generates digital samples of two sine waves precisely offset by 90 degrees in phase creating sine and cosine signals [8], [10], [11], (See Figure 2). It uses a digital phase accumulator (adder) and sine/cosine look-up tables. The digital samples out of the local oscillator are generated at a sampling rate exactly equal to the ADC sample clock frequency, fs. Since the data rates from these two mixer input sources are both at the ADC sampling rate, fs, the complex mixer output samples at fs. http://www.iosrjournals.org/iosr-jece/papers/Vol.%2011%20Issue%206/Version-4/B1106040513.pdf
milstar: https://radioprog.ru/post/415
milstar: http://ptgmedia.pearsoncmg.com/images/9780137027415/samplepages/0137027419.pdf Richard Lyons is a consulting systems engineer and lecturer with Besser Associates in Mountain View, California. He has been the lead hardware engineer for numerous signal processing systems for both the National Security Agency (NSA) and Northrop Grumman Corp.
milstar: https://www.analog.com/media/en/technical-documentation/data-sheets/AD6620.pdf DDC 21$ 14 bit not for new design https://www.renesas.com/eu/en/www/doc/datasheet/hsp50016.pdf https://www.renesas.com/eu/en/www/doc/application-note/an9401.pdf DDC 16 bit 75 msps
milstar: 2.5.3. Ñèíõðîííîå äåòåêòèðîâàíèå https://ozlib.com/831358/tehnika/sinhronnoe_detektirovanie
milstar: https://www.analog.com/media/en/technical-documentation/data-sheets/AD6636.pdf
milstar: http://www.geo.uzh.ch/microsite/rsl-documents/research/SARlab/GMTILiterature/PDF/Skolnik90.pdf
milstar: As mentioned in the first page of this chapter, How to Demodulate an AM Waveform, one approach to amplitude demodulation involves multiplying the received signal by a carrier-frequency reference signal, and then low-pass-filtering the result of this multiplication. This method provides higher performance than AM demodulation that is built around a leaky peak detector. However, this approach has a serious weakness: the result of the multiplication is affected by the phase relationship between the transmitter’s carrier and the receiver’s carrier-frequency reference signal. These plots show the demodulated signal for three values of transmitter-to-receiver phase difference. As the phase difference increases, the amplitude of the demodulated signal decreases. The demodulation procedure has become nonfunctional at 90° phase difference; this represents the worst-case scenario—i.e., the amplitude begins to increase again as the phase difference moves away (in either direction) from 90° One way to remedy this situation is through additional circuitry that synchronizes the phase of the receiver’s reference signal with the phase of the received signal. However, quadrature demodulation can be used to overcome the absence of synchronization between transmitter and receiver. As was just pointed out, the worst-case phase discrepancy is ±90°. Thus, if we perform multiplication with two reference signals separated by 90° of phase, the output from one multiplier compensates for the decreasing amplitude of the output from the other multiplier. In this scenario the worst-case phase difference is 45°, and you can see in the above plot that a 45° phase difference does not result in a catastrophic reduction in the amplitude of the demodulated signal. =============================================================================================================================== The following plots demonstrate this I/Q compensation. The traces are demodulated signals from the I and Q branches of a quadrature demodulator. Transmitter phase = 0° Transmitter phase = 45° (the orange trace is behind the blue trace—i.e., the two signals are identical) Transmitter phase = 90° https://www.allaboutcircuits.com/textbook/radio-frequency-analysis-design/radio-frequency-demodulation/understanding-quadrature-demodulation/ Constant Amplitude It would be convenient if we could combine the I and Q versions of the demodulated signal into one waveform that maintains a constant amplitude regardless of the phase relationship between transmitter and receiver. Your first instinct might be to use addition, but unfortunately it’s not that simple. The following plot was generated by repeating a simulation in which everything is the same except the phase of the transmitter’s carrier. The phase value is assigned to a parameter that has seven distinct values: 0°, 30°, 60°, 90°, 120°, 150°, and 180°. The trace is the sum of the demodulated I waveform and the demodulated Q waveform. As you can see, addition is certainly not the way to produce a signal that is not affected by variations in the transmitter-to-receiver phase relationship. This is not surprising if we remember the mathematical equivalence between I/Q signaling and complex numbers: the I and Q components of a signal are analogous to the real and imaginary parts of a complex number. By performing quadrature demodulation, we obtain real and imaginary components that correspond to the magnitude and phase of the baseband signal. In other words, I/Q demodulation is essentially translation: we are translating from a magnitude-plus-phase system (used by a typical baseband waveform) to a Cartesian system in which the I component is plotted on the x-axis and the Q component is plotted on the y-axis. To obtain the magnitude of a complex number, we can’t simply add the real and imaginary parts, and the same applies to I and Q signal components. Instead, we have to use the formula shown in the diagram, which is nothing more than the standard Pythagorean approach to finding the length of the hypotenuse of a right triangle. ============================================================ If we apply this formula to the I and Q demodulated waveforms, we can obtain a final demodulated signal that is not affected by phase variations. The following plot confirms this: the simulation is the same as the previous one (i.e., seven different phase values), but you see only one signal, because all the traces are identical. https://www.allaboutcircuits.com/textbook/radio-frequency-analysis-design/radio-frequency-demodulation/how-to-demodulate-an-am-waveform/
milstar: http://www.ni.com/tutorial/4805/en/ https://www.keysight.com/upload/cmc_upload/All/6-Analysis-of-Baseband.pdf I = 1.059*cos(2π ∗0.624MHz ∗t) Q = −sin(2π ∗0.624MHz ∗t)
milstar: http://www.farnell.com/datasheets/25526.pdf https://www.analog.com/media/en/technical-documentation/application-notes/AN-924.pdf
milstar: https://areeweb.polito.it/didattica/corsiddc/01NVD/ATLCE10/Lessons/ATLCEC51.pdf
milstar: https://datasheets.maximintegrated.com/en/ds/MAX2022.pdf The MAX2022 utilizes an internal passive mixer architecture. This enables a very low noise floor of -173.2dBm/Hz for low-level signals, below about -20dBm output power level. For higher output level signals, the noise floor will be determined by the internal LO noise level at approximately -162dBc/Hz. https://datasheets.maximintegrated.com/en/ds/MAX2023.pdf https://datasheets.maximintegrated.com/en/ds/MAX2021.pdf ♦ 0.06dB Typical I/Q Gain Imbalance ♦ 0.15° I/Q Typical Phase Imbalance ♦ +35.2dBm Typical IIP3 ♦ +76dBm Typical IIP2 ♦ > 30dBm IP1dB ♦ 9.2dB Typical Conversion Loss ♦ 9.3dB Typical NF
milstar: https://www.jlab.org/intralab/calendar/archive01/LLRF/ziomek.pdf
milstar: http://www.ti.com/lit/ds/symlink/trf371125.pdf
milstar: https://indico.desy.de/indico/event/3391/session/5/contribution/97/material/slides/0.pdf IQ sampling Fclock=4FS
milstar: http://airspot.ru/book/file/961/radar_handbook.pdf 2008
milstar: https://mydocx.ru/10-107285.html
milstar: https://apps.dtic.mil/dtic/tr/fulltext/u2/a264466.pdf
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