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Operazionnie ysiliteli ,ZAP/AZP & (ïðîäîëæåíèå)
milstar: 1941: First (vacuum tube) op-amp An op-amp, defined as a general-purpose, DC-coupled, high gain, inverting feedback amplifier, is first found in US Patent 2,401,779 "Summing Amplifier" filed by Karl D. Swartzel Jr. of Bell labs in 1941. This design used three vacuum tubes to achieve a gain of 90dB and operated on voltage rails of ±350V. ###################################################### It had a single inverting input rather than differential inverting and non-inverting inputs, as are common in today's op-amps. Throughout World War II, Swartzel's design proved its value by being liberally used in the M9 artillery director designed at Bell Labs. ######################################################################### This artillery director worked with the SCR584 radar system to achieve extraordinary hit rates (near 90%) that ####################################################################### would not have been possible otherwise.[3] ########################### http://en.wikipedia.org/wiki/Operational_amplifier
milstar: The new IOADCDACD1.5G mezzanine card from Annapolis Micro Systems was built specifically for use as a DRFM, and it has some impressive specs. Its two analog input and output channels have around 600 MHz instantaneous bandwidth each, which is respectable and covers a good range of applications. The board’s latency, though, is where it really shines. The latency from SMA to SMA connector is only 39 ns when going through the user FPGA space, or even lower when using a digital bypass. This makes it well suited to self-protection applications. Its board support package is also designed to make the integration of a user’s DRFM kernel simple, which should be refreshing to anyone who has had to build a DRFM out of an FPGA board that wasn’t really designed to be one” – a DRFM Industry Expert The WILDSTAR G2 Dual 1.5 GSps 12-Bit ADC & DAC Mezzanine Card is shipped with a custom heatsink which enables proper cooling of the ADC. An on-board temperature monitor is also supplied which allows for real-time monitoring of the ADC’s internal die temperature. The WILDSTAR G2 Dual 1.5 GSps 12-Bit ADC & DAC Mezzanine Card provides high fidelity and high speed analog-to-digital conversion along with a rugged design. https://www.annapmicro.com/products/wildstar-dual-1-5-gsps-12-bit-adcdac-converter-mezzanine-card/
milstar: Posted on June 5, 2014 by aluongo Annapolis, Maryland – June 5, 2014 – Annapolis Micro Systems, Inc. announced today that they have been recognized by Lockheed Martin Corporation as one of their Outstanding Suppliers. Though Lockheed Martin has thousands of suppliers, only a few small businesses are recognized for doing an outstanding job of providing products and support during the last year and for their outstanding quality in goods and services. This entry was posted in Featured News, Press Releases. Bookmark the permalink. Post navigation
milstar: https://prezi.com/tesgt-dwnlkj/digital-radio-frequency-memory-drfm/ DRFM sozdaet dlja radara loznuju cel
milstar: file:///C:/Users/Mubariz/Downloads/Advanced-SIGINT-Capability-for-a-SWaP-constrained-Platform-case-study.pdf
milstar: Common SIGINT System 4000 http://www.northropgrumman.com/Capabilities/AirborneSIGINTProductLine/Documents/CSS-4000_datasht.pdf
milstar: http://www.e2v-us.com/shared/content/resources/File/documents/broadband-data-converters/doc0869B.pdf
milstar: EV12AS200VZPY http://www.e2v.com/content/uploads/2014/09/EV12AS200xZPY-Qualification-Report.pdf -40 +110 grad C Die size 4.82 mm * 4.82 mm Wafer lab - Infineon FpBGA 196 15*15*1.2 mm 200 Ghz SiGE Bipolar Pitch -1 mm
milstar: EV12AS200ZPY http://www.e2v.com/resources/account/download-datasheet/1784 Gain flatness 1300 -1800 mhz 1.33 gaps -0.9db SINAD 1490 mhz 1.5 gsps 55/52 dBFS ENOB -8.8/8.4 bit SFDR 67/55 dBFs
milstar: EV12AS200ZPY http://www.e2v.com/resources/account/download-datasheet/1784 Gain flatness 1300 -1800 mhz 1.33 gaps -0.9db SINAD 1490 mhz 1.5 gsps 55/52 dBFS ENOB -8.8/8.4 bit SFDR 67/55 dBFs
milstar: http://www.apissys.com/views/media_produit/datasheets/11/AF202-0.pdf 2 *1.5 gsps 12 bit Abc -50 gramm Fin 1.5 ghz, snd -55 dBFs ,SFDR -67dbc,ENOB -8.7 bit 2*15 gsps
milstar: Why Oversample when Undersampling can do the Job? System designers most often tend to use ADC sampling frequency as twice the input signal frequency. As an example, for a signal with 70-MHz input signal frequency with 20-MHz signal bandwidth, system designers often use more than 140 MSPS sampling rate for ADC even though anything above 40 MSPS is sufficient as the sampling rate. http://www.ti.com/lit/an/slaa594a/slaa594a.pdf Nyquist-Shannon Sampling theorem, which is the modified version of the Nyquist sampling theorem, says that the sampling frequency needs to be twice the signal bandwidth and not twice the maximum frequency component, in order to be able to reconstruct the original signal perfectly from the sampled version. If B is the signal bandwidth, then Fs > 2B is required where Fs is sampling frequency. The signal bandwidth can be from DC to B or from f1 to f2 where B = f2 – f1. The aliasing effect due to the undersampling technique can be used for our advantage. When a signal is sampled at a rate less than twice its maximum frequency, the aliased signal appears at Fs – Fin, where Fs is the sampling frequency and Fin in the input signal frequency. In the above case, if we sample the 70-MHz signal with 100 MSPS sampling rate, the aliased component will appear at 30 MHz (100 – 70). As we know in advance that the signal is aliased, we can recover the actual frequency by using the Fs – Fin relationship. The undersampling technique allows the ADC to behave like a mixer or a down converter in the receive chain. For a band-limited signal of 70 MHz with a 20-MHz signal bandwidth, if the sampling rate (Fs) is 100 MSPS, the aliased component will appear between 20 MHz to 40 MHz (30 ±10 MHz). We will use the example of a 70-MHz signal with 20-MHz bandwidth (60 MHz to 80 MHz) for the discussion throughout this paper. For a radar application and for communication systems, generally 70 MHz is used as IF (intermediate frequency) with a specific bandwidth ranging from a few KHz to a few MHz. The maximum frequency component is 80 MHz in this signal. For an oversampling case, the minimum sampling rate is more than 160 MSPS. To keep this band of 60 MHz to 80 MHz in the middle of the first Nyquist Zone, the sampling frequency is 280 MSPS. This signal in frequency domain is shown in Figure 1.
milstar: Are there any Advantages of Oversampling? Yes. There are some specific advantages of oversampling which are described below. 3.1 Processing Gain When the signal is oversampled a greater number of times than its signal bandwidth, then the processing gain is achieved in addition to the SNR shown in the ADC datasheets. For example, for the ADS4149, at 70 MHz, the SNR will be around 72 dB at the sampling rate of 200 MSPS. For our example of 70 MHz with 20-MHz bandwidth, the signal is oversampled by 10 times with respect to signal bandwidth. Note that with respect to the signal frequency of 70 MHz, it is oversampled only around 3 times. Due to oversampling of 10 times to the bandwidth, system designers get the extra advantage of processing gain in addition to the actual SNR mentioned in the datasheet. For Fs of 200 MSPS, the SNR of 72 dB is for a Nyquist bandwidth of Fs/2, that is, 100 MHz. For the measurement of SNR of the ADC, the noise in the entire band of 100 MHz is considered in this case. The processing gain is achieved by using the following formula: Process Gain = 10 log ((Fs/2)/BW) Where Fs is the sampling Rate; BW is the signal bandwidth; For the oversampling example, BW is 20 MHz, Fs is 200 MHz. If we use the above formula, the processing gain is around 7dB. The total SNR can be calculated using the following formula: SNRtotal = SNRds + Process Gain Where SNRtotal is the total SNR after adding the processing gain and SNRds is the SNR value provided in the datasheet (without the processing gain). SNRtotal is 79 dBFS (72 + 7) using the above formula.
milstar: ADS54J60 Dual-Channel, 16-Bit, 1.0-GSPS Analog-to-Digital Converter http://www.ti.com/lit/ds/symlink/ads54j60.pdf SNR Fin 170 mhz -70dBFS 69.8 dBFS, SFDR -88 dbc Fin 350 mhz -67.5 dBFS Input 1.9 pp ---------------- Dlja Fin 170 mhz BW= 20 mhz Process Gain = 10 log ((Fs/2)/BW) 10 log (500/20)=25 = +14 db 69.8 +14 db = 83.8 dBFS SINAD dlja BW =40 mhz 80.8 dBFS The ADS54J60 has a thermal noise of approximately 71.1 dBFS and an internal aperture jitter of 145 fs. The SNR, depending on the amount of external jitter for different input frequencies, is shown in Figure 124.
milstar: For example, consider radar using a 30-MHz bandwidth waveform at an IF of 800 MHz. If this is sampled using an A/D converter at a sample rate of 2.0 GSPS to a resolution of 12 bits, the output bandwidth of the data would be 1000 MHz, far in excess of the signal bandwidth, and the output data rate from the converter would be 3.0 GBytes/s. http://www.edn.com/design/analog/4431036/3/Demand-for-digital--Challenges-and-solutions-for-high-speed-ADCs-and-RADAR-systems If the data is decimated by a factor of 16 using a DDC, not only does the decimation provide some increased noise reduction but the output data rate is reduced to below 625 MBytes/s, which enables data transportation using only a single JESD204B lane! This significantly reduces the overall system power required. With the ability to dynamically configure the DDCs or bypass them as needed, new high-speed ADCs provide the option of switching between different modes to support power and implement optimized solutions as needed and enable the feature sets needed for cognitive radar applications. New GSPS ADCs provide solutions to not only overcome existing challenges but to further optimize the system. In supporting digitization closer to the antenna these converters provide unparalleled linearity as well as an analog bandwidth of over 3 GHz, enabling under-sampling of the L and most of the S frequency bands. This enables direct RF sampling within these frequency bands, reducing component count and system size by eliminating mixer stages. For higher frequency systems this also enables higher IFs to be used, providing options for reducing the number of mixing stages and filters, as well as increasing the frequency planning options as a wide range of IFs can be used. ------------- The latest GSPS ADCs are able to provide in excess of 75 dBc SFDR, which is nearly a 20 dBc improvement over devices that have been available in the last decade. This significant leap is even more critical when competing with recent communications infrastructure frequency allocation To explore this further, Figure 1 illustrates a high level overview of a typical current X-band radar system. Within this system two analog mixing stages are typically utilized. The first stage mixes the pulsed radar return to a frequency of around 1 GHz and the second to an IF in the region of 100 to 200 MHz to enable sampling of the signal using a 200 MSPS or lower A/D converter, to a resolution of 12 bits or higher. --------------------- Using a GSPS converter with an analog bandwidth in excess of 1.5 GHz already supports digitization of the first IF, but in many cases the performance of current GSPS ADCs has limited the acceptability of this solution as the linearity and noise spectral density of the device has not met the system requirements ---------------
milstar: As with modern test equipment, such as spectrum analyzers—many of which rely on high-speed ADCs and digital processing following an input signal path with frequency downconversion to an IF section—modern radar systems are as much defined by their digital circuitry as by their analog RF/microwave circuitry. The bandwidth and sampling rates of the ADCs set the limits for the radar’s IF stage, while the resolution of the ADCs (in bits) determines the resolution of the radar system receiver. Similarly, the DACs help generate complex modulated pulsed signals for a radar transmitter, relying on frequency upconversion and trusted RF/microwave components (such as amplifiers and filters) for the signal path to the radar system’s transmit antennas. http://defenseelectronicsmag.com/systems-amp-subsystems/radar-systems-now-rely-data-converters
milstar: The ADC used is ADC12b1800RF from Texas Instruments. It is based on calibrated folding and interpolating architecture and has an input bandwidth of 2.7 GHz in non-DES mode and 1.2 GHz in DESI and DESQ mode, maximum sampling rate of 3.6 GSPS in interleaved mode and 1800 MSPS in dual ADC mode, 12 bit of output. ENOB of 8.7 bits, SNR of 54.3 dB, SFDR of 64 dBC for I/P = 1448 MHz @ -0.5 dBFS. Linear Frequency Modulated (LFM) waveform with 10us pulse width and center frequency of 1.3 GHz http://www.radarindia.com/irsi13papers/13-FP-122.pdf
milstar: Noise Considerations in High Speed Converter Signal Chains http://www.analog.com/media/en/training-seminars/tutorials/MT-230.pdf
milstar: http://www.gdsatcom.com/Electronics/Data%20Sheets/14368_C.pdf X-Band Low Noise Amplifiers LXA-7500 Series
milstar: http://www.digikey.com/web%20export/supplier%20content/TI_296/mkt/imaging/radar.pdf?redirected=1
milstar: Overview of Radar DMTI Processing The SPS-48E radar (Fig. 1) uses a triple conversion receiver. ########### The system is wideband until the second intermediate frequency (IF) conversion, where the individual beams are bandpass filtered and separated. Since three beams are used in the DMTI, there are three coherent oscillator frequencies (one for each beam) in the final conversion of the receiver (final IF is about 1.5 MHz). ################ A single analog-to-digital (A/D) converter is used for each beam. In-phase and quadrature (I/Q) data are developed based on samples that are spaced at multiples of 90° at the IF frequency. The interpolation filter develops the I/Q estimates from A/D samples (see the boxed insert, Intermediate-Frequency Sampling Technique). The I/Q data preserve the amplitude and phase of the IF radar return. The amplitude of the radar return is computed as ( ). I Q 2 2 + The phase of the return is computed as tan21 (Q/I). From pulse to pulse, a phase progression will be seen on moving targets due to Doppler, and no phase progression will be seen on stationary reflectors. It is this phase progression on moving targets that allows such targets to be separated from stationary reflectors (clutter). To remove clutter and pass targets, DMTI filters are employed in each beam independently. A bank of digital filters is used to cover the region between low velocity (small phase shift per pulse) and higher velocity (near 360° phase shift per pulse). Targets moving at speeds such that they present more than a 360° phase progression per pulse are said to be velocity ambiguous, since the radar pulse repetition interval causes aliasing. For example, a phase progression of 400° per pulse appears exactly as a phase progression of 40° per pulse. To avoid velocity blinds (i.e., targets moving at speeds such that their phase progression is 360° per pulse, thus appearing as 0° per pulse), the pulse repetition frequency is jittered on a burst-to-burst basis. This ensures that the phase progression presented by the target will vary on a burst-to-burst basis, and thus the target will not be velocity blinded on all bursts http://www.jhuapl.edu/techdigest/TD/td1803/roul.pdf Description The AN/SPS-48G is a long-range, three-dimensional (3D) Air Search Radar that will be installed on CVN, LHA, LHD, and LPD 17 class ships. The AN/SPS-48G is used to find full volumetric detection data for Ships Self Defense System and the Cooperative Engagement Capability (CEC), Air Intercept Control, Anti-Ship Cruise Missile detection including Low Elevation and High Diver targets, backup aircraft marshalling, and the new Hazardous Weather Detection and Display Capability. http://www.navy.mil/navydata/fact_display.asp?cid=2100&tid=1250&ct=2 AN/SPS-48E - Compared to the C variant, the SPS-48E has twice the radiated power, increased receiver sensitivity, four stage solid-state transmitter, half the components of a -48C and built-in testing for easier diagnostics. Originally developed as part of the New Threat Upgrade (NTU) Program to support the SM-2 Launch On Search (LOS) capability. 1975 under the Guided Missile Frigate Anti-Air Warfare Modernization Program. The AN/SPS-48E included a digital receiver and signal processor that could automatically detect and track very small targets, even when jammed. It was included in the New Threat Upgrade of the 1980s. The deployment of the AN/SPY-1 and the end of the Cold War led to the decommissioning of a large number of such ships, and many of these vessels AN/SPS-48 sets were reused on aircraft carriers and amphibious ships, where it is used to direct targets for air defense systems such as the Sea Sparrow and RIM-116 SAM missiles. Existing sets are being modernized under the ROAR program to AN/SPS-48G standard for better reliability and usability. ################# INTERMEDIATE-FREQUENCY SAMPLING TECHNIQUE To develop in-phase (I) and quadrature (Q) data, the SPS-48E radar uses an intermediate-frequency (IF) sampling technique with an IF bandwidth of approximately 400 kHz, IF center frequency of about 1.5 MHz, and analog-to-digital (A/D) sampling frequency of 6 MHz. There is a precise 4:1 relation between the IF sample frequency and the IF center frequency. If modulation effects across the received pulsewidth are ignored, the echo may be thought of as several cycles of a sine wave. The sine wave is sampled at four times its rate, i.e., every 90°. Therefore, alternate samples will be in quadrature with each other. To account for modulation effects across the pulse, one sample is defined to be “I”; two leading and two trailing samples are combined by the following equation to create the “Q” sample (s): 180° phase shift 90° phase shift Time Q ssss =− − + + 1 16 9 16 9 16 1 16 1234 Q I QI Q I Q s s Is s −− −− 1234 This technique provides accuracy acceptable for the clutter cancellation requirements of the SPS-48E lowelevation-mode DMTI. If higher clutter cancellation is required, a more elaborate finite impulse response filter for both the I and the Q channel is required. The advantage of the current technique is that I/Q data are developed with only a single A/D converter. The two baseband analog channels in a conventional receiver are not required, and aliasing due to channel gain mismatch is avoided. Amplitude modulation effects across the received pulse do, however, cause some degradation.
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